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622 mbps clock and data recovery ic with integrated limiting amplifier adn2804 rev. b information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. specifications subject to change without notice. no license is granted by implication or otherwise under any patent or patent rights of analog devices. trademarks and registered trademarks are the property of their respective owners. one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 781.329.4700 www.analog.com fax: 781.461.3113 ?2006C2010 analog devices, inc. all rights reserved. features exceeds sonet requirements for jitter transfer/ generation/tolerance quantizer sensitivity: 3.3 mv typical adjustable slice level: 95 mv patented clock recovery architecture loss-of-signal (los) detect range: 2.6 mv to 18.4 mv independent slice level adjust and los detector no reference clock required loss-of-lock indicator i 2 c? interface to access optional features single-supply operation: 3.3 v low power: 423 mw typical 5 mm 5 mm, 32-lead lfcsp, pb free applications bpon ont sonet oc-12 wdm transponders regenerators/repeaters test equipment broadband cross-connects and routers general description the adn2804 provides the receiver functions of quantization, signal level detect, clock and data recovery, and data retiming for 622 mbps nrz data. the adn2804 automatically locks to 622 mbps data without the need for an external reference clock or programming. in the absence of input data, the output clock drifts no more than 5%. all sonet jitter requirements are met, including jitter transfer, jitter generation, and jitter tolerance. all specifications are quoted for ?40c to +85c ambient temperature, unless otherwise noted. this device, together with a pin diode and a tia preamplifier, can implement a highly integrated, low cost, low power fiber optic receiver. the receivers front-end loss-of-signal (los) detector circuit indicates when the input signal level falls below a user-adjustable threshold. the los detect circuit has hysteresis to prevent chatter at the output. the adn2804 is available in a compact 5 mm 5 mm, 32-lead lfcsp. functional block diagram 2 slicep/slicen lol dataoutp/ dataoutn los thradj clkoutp/ clkoutn adn2804 2 vcc vee cf1 cf2 pin nin vref quantizer vco phase shifter phase detect frequency detect los detect data re-timing loop filter loop filter refclkp/refclkn (optional) 2 0 5801-001 figure 1.
adn2804 rev. b | page 2 of 24 table of contents features .............................................................................................. 1 applications ....................................................................................... 1 general description ......................................................................... 1 functional block diagram .............................................................. 1 revision history ............................................................................... 2 specifications ..................................................................................... 3 jitter specifications ....................................................................... 4 output and timing specifications ............................................. 5 absolute maximum ratings ............................................................ 6 thermal characteristics .............................................................. 6 esd caution .................................................................................. 6 timing characteristics ..................................................................... 7 pin configuration and function descriptions ............................. 8 typical performance characteristics ............................................. 9 i 2 c interface timing and internal register description ........... 10 terminology .................................................................................... 12 jitter specifications ......................................................................... 13 theory of operation ...................................................................... 14 functional description .................................................................. 16 frequency acquisition ............................................................... 16 limiting amplifier ..................................................................... 16 slice adjust .................................................................................. 16 loss-of-signal (los) detector ................................................. 16 lock detector operation .......................................................... 17 squelch modes ...................................................................... 17 i 2 c interface ................................................................................ 17 reference clock (optional) ...................................................... 19 applications information .............................................................. 21 pcb design guidelines ............................................................. 21 dc-coupled application .......................................................... 23 outline dimensions ....................................................................... 24 ordering guide .......................................................................... 24 revision history 5/10rev. a to rev. b changes to figure 5 and table 5 ..................................................... 8 changes to figure 24 ...................................................................... 21 2/09rev. 0 to rev. a updated outline dimensions ....................................................... 24 changes to ordering guide .......................................................... 24 2/06revision 0: initial version
adn2804 rev. b | page 3 of 24 specifications t a = t min to t max , vcc = v min to v max , vee = 0 v, c f = 0.47 f, slicep = slicen = vee, input data pattern: prbs 2 23 ? 1, unless otherwise noted. table 1. parameter conditions min typ max unit quantizerdc characteristics input voltage range @ pin or nin, dc-coupled 1.8 2.8 v peak-to-peak differential input pin ? nin 2.0 v input common-mode level dc-coupled (see figure 27 , figure 28 , and figure 29 ) 2.3 2.5 2.8 v differential input sensitivity 2 23 ? 1 prbs, ac-coupled, 1 ber = 1 10 C10 6 3.3 mv p-p input offset 500 v input rms noise ber = 1 10 C10 290 v rms quantizerac characteristics data rate 622 mbps output clock range absence of input data 622 5% mhz s11 @ 622 mhz ?15 db input resistance differential 100 input capacitance 0.65 pf quantizerslice adjustment gain slicep ? slicen = 0.5 v 0.10 0.11 0.13 v/v differential control voltage input slicep ? slicen ?0.95 +0.95 v control voltage range dc level @ slicep or slicen vee 0.95 v slice threshold offset 1 mv loss-of-signal (los) detect loss-of-signal detect range (see figure 6 ) r thresh = 0 14.9 16.7 18.4 mv r thresh = 100 k 2.6 3.5 4.4 mv hysteresis (electrical) oc-12 r thresh = 0 6.2 6.9 7.7 db r thresh = 100 k 4.1 6.1 8.1 db los assert time dc-coupled 2 500 ns los deassert time dc-coupled 2 400 ns loss-of-lock (lol) detect vco frequency error for lol assert with respect to nominal 1000 ppm vco frequency error for lol deassert with respect to nominal 250 ppm lol response time oc-12 200 s acquisition time lock to data mode oc-12 2.0 ms optional lock to refclk mode 20.0 ms data rate readback accuracy fine readback in addition to refclk accuracy oc-12 100 ppm power supply voltage 3.0 3.3 3.6 v power supply current locked to 622.08 mbps 128 ma operating temperature range C40 +85 c 1 pin and nin should be differentially driven and ac-coupled for optimum sensitivity. 2 when ac-coupled, the los assert and deasse rt times are dominated by the rc time constant of the ac coupling capacitor and the 50 input termination of the adn2804 input stage.
adn2804 rev. b | page 4 of 24 jitter specifications t a = t min to t max , vcc = v min to v max , vee = 0 v, c f = 0.47 f, slicep = slicen = vee, input data pattern: prbs 2 23 ? 1, unless otherwise noted. table 2. parameter conditions min typ max unit phase-locked loop characteristics jitter transfer bandwidth oc-12 75 130 khz jitter peaking oc-12 0 0.03 db jitter generation oc-12, 12 khz to 5 mhz 0.001 0.003 ui rms 0.011 0.026 ui p-p jitter tolerance oc-12, 2 23 ? 1 prbs 30 hz 1 100 ui p-p 300 hz 1 44 ui p-p 25 khz 2.5 ui p-p 250 khz 1 1.0 ui p-p 1 jitter tolerance of the adn2804 at these jitter frequencies is better than what the test equipm ent is able to measure.
adn2804 rev. b | page 5 of 24 output and timing specifications table 3. parameter conditions min typ max unit lvds output characteristics (clkoutp/clkoutn, dataoutp/dataoutn) output voltage high v oh (see figure 3 ) 1475 mv output voltage low v ol (see figure 3 ) 925 mv differential output swing v od (see figure 3 ) 250 320 400 mv output offset voltage v os (see figure 3 ) 1125 1200 1275 mv output impedance differential 100 lvds outputs timing rise time 20% to 80% 115 220 ps fall time 80% to 20% 115 220 ps setup time t s (see figure 2 ), oc-12 760 800 840 ps hold time t h (see figure 2 ), oc-12 760 800 840 ps i 2 c interface dc characteristics lvcmos input high voltage v ih 0.7 vcc v input low voltage v il 0.3 vcc v input current v in = 0.1 vcc or v in = 0.9 vcc ?10.0 +10.0 a output low voltage v ol , i ol = 3.0 ma 0.4 v i 2 c interface timing see figure 11 sck clock frequency 400 khz sck pulse width high t high 600 ns sck pulse width low t low 1300 ns start condition hold time t hd;sta 600 ns start condition setup time t su;sta 600 ns data setup time t su;dat 100 ns data hold time t hd;dat 300 ns sck/sda rise/fall time t r /t f 20 + 0.1 cb 1 300 ns stop condition setup time t su;sto 600 ns bus free time between a stop and a start t buf 1300 ns refclk characteristics optional lock to refclk mode input voltage range @ refclkp or refclkn v il 0 v v ih vcc v minimum differential input drive 100 mv p-p reference frequency 10 160 mhz required accuracy 100 ppm lvttl dc input characteristics input high voltage v ih 2.0 v input low voltage v il 0.8 v input high current i ih , v in = 2.4 v 5 a input low current i il , v in = 0.4 v ?5 a lvttl dc output characteristics output high voltage v oh , i oh = ?2.0 ma 2.4 v output low voltage v ol , i ol = +2.0 ma 0.4 v 1 c b = total capacitance of one bus line in picofarads. if used with hs-mode devices, faster fall times are allowed.
adn2804 rev. b | page 6 of 24 absolute maximum ratings t a = t min to t max , vcc = v min to v max , vee = 0 v, c f = 0.47 f, slicep = slicen = vee, unless otherwise noted. table 4. parameter rating stress above those listed under absolute maximum ratings may cause permanent damage to the device. this is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. supply voltage (vcc) 4.2 v minimum input voltage (all inputs) vee ? 0.4 v maximum input voltage (all inputs) vcc + 0.4 v maximum junction temperature 125c storage temperature range ?65c to +150c thermal characteristics thermal resistance 32-lead lfcsp, 4-layer board with exposed paddle soldered to vee, ja = 28c/w. esd caution esd (electrostatic discharge) sensitive device. electros tatic charges as high as 4000 v readily accumulate on the human body and test equipment and can discharge wi thout detection. although this product features proprietary esd protection circuitry, permanent dama ge may occur on devices subjected to high energy electrostatic discharges. therefore, proper esd precautions are recommended to avoid performance degradation or loss of functionality.
adn2804 rev. b | page 7 of 24 timing characteristics clkoutp dataoutp/ dataoutn t s t h 05801-002 figure 2. output timing |v od | v oh differential clkoutp/n, d a t a outp/n v os v ol 0 5801-032 figure 3. differential output specifications simplified lvds output stage r load 100? 100? 5ma 5ma v diff 05801-033 figure 4. differential output stage
adn2804 rev. b | page 8 of 24 pin configuration and fu nction descriptions 05801-004 vcc1 vcc 2 vref 3 pin 1 indic ator top view (not to scale) 24 vcc 23 vee 22 los 21 sda 32 vcc 20 sck 19 saddr5 18 vcc 17 vee thradj 9 refclkp 10 refclkn 11 vcc 12 vee 13 cf2 14 cf1 15 lol 16 nin 4 pin 5 slicep 6 slicen 7 vee 8 31 vcc 30 vee 29 dataoutp 28 dataoutn 27 squelch 26 clkoutp 25 clkoutn adn2804* * there is an exposed pad on the bottom of the package that must be connected to gnd. figure 5. pin configuration table 5. pin function descriptions pin no. mnemonic type 1 description 1 vcc ai connect to vcc. 2 vcc p power for limiting amplifier, los. 3 vref ao internal vref voltage. deco uple to gnd with a 0.1 f capacitor. 4 nin ai differential data input. cml. 5 pin ai differential data input. cml. 6 slicep ai differential slice level adjust input. 7 slicen ai differential slice level adjust input. 8 vee p gnd for limiting amplifier, los. 9 thradj ai los threshold setting resistor. 10 refclkp di differential refclk input. 10 mhz to 160 mhz. 11 refclkn di differential refclk input. 10 mhz to 160 mhz. 12 vcc p vco power. 13 vee p vco gnd. 14 cf2 ao frequency loop capacitor. 15 cf1 ao frequency loop capacitor. 16 lol do loss-of-lock indicator. lvttl active high. 17 vee p fll detector gnd. 18 vcc p fll detector power. 19 saddr5 di slave address bit 5. 20 sck di i 2 c clock input. 21 sda di i 2 c data input. 22 los do loss-of-signal detect output. active high. lvttl. 23 vee p output buffer, i 2 c gnd. 24 vcc p output buffer, i 2 c power. 25 clkoutn do differential recovered clock output. lvds. 26 clkoutp do differential recovered clock output. lvds. 27 squelch di disable clock and data outputs. active high. lvttl. 28 dataoutn do differential recovered data output. lvds. 29 dataoutp do differential recovered data output. lvds. 30 vee p phase detector, phase shifter gnd. 31 vcc p phase detector, phase shifter power. 32 vcc ai connect to vcc. exposed pad pad p connect to gnd. works as a heat sink. 1 type: p = power, ai = analog input, ao = analog output, di = digital input, do = digital output.
adn2804 rev. b | page 9 of 24 typical performance characteristics 16 14 12 1 10 100 1k 10k 100k 10 8 6 4 2 trip point (mv p-p) 0 5801-005 figure 6. los comparator trip point programming
adn2804 rev. b | page 10 of 24 i 2 c interface timing and intern al register description 1a500000x msb = 1 set by pin 19 0 = wr 1 = rd slave address [6 ...0] r/w ctrl. 0 5801-007 figure 7. slave address configuration s slave addr, lsb = 0 (wr) a(s) a(s) a(s) data sub addr a(s) p data 0 5801-00 8 figure 8. i 2 c write data transfer s s = start bit p = stop bit a(s) = acknowledge by slave a(m) = acknowledge by master a(m) = lack of acknowledge by master s slave addr, lsb = 0 (wr) slave addr, lsb = 1 (rd) a(s) a(s) sub addr a(s) data a(m) data p a(m) 0 5801-009 figure 9. i 2 c read data transfer start bit s stop bit p ack ack wr ack d0 d7 a0 a7 a5 a6 sladdr[4...0] slave address sub address data sub addr[6...1] data[6...1] sck sda 0 5801-010 figure 10. i 2 c data transfer timing t buf sda t f t t r t f t su;dat t su;sto t hd;sta t r ss p s sck low t hd;sta t hd;dat t high t su;sta 0 5801-011 figure 11. i 2 c port timing diagram
adn2804 rev. b | page 11 of 24 table 6. internal register map 1 reg name r/w addr d7 d6 d5 d4 d3 d2 d1 d0 freq0 r 0x0 msb lsb freq1 r 0x1 msb lsb freq2 r 0x2 0 msb lsb misc r 0x4 x x los status static lol lol status data rate measurement complete x x ctrla w 0x8 f ref range data rate/div_f ref ratio measure data rate lock to reference ctrlb w 0x9 config lol reset misc[4] system reset 0 reset misc[2] 0 0 0 ctrlc w 0x11 0 0 0 0 0 config los squelch mode output boost 1 all writeable registers default to 0x00. table 7. miscellaneous register, misc los status static lol lol status data rate measurement complete d7 d6 d5 d4 d3 d2 d1 d0 x x 0 = no loss of signal 0 = waiting for next lol 0 = locked 0 = measuring data rate x x 1 = loss of signal 1 = static lol until reset 1 = acquiring 1 = measurement complete table 8. control register, ctrla 1 f ref range data rate/div_f ref ratio measure data rate lock to reference d7 d6 d5 d4 d3 d2 d1 d0 0 0 19.44 mhz 0 1 0 1 32 set to 1 to measure data rate 0 = lock to input data 0 1 38.88 mhz 0 1 0 1 32 1 = lock to reference clock 1 0 77.76 mhz 0 1 0 1 32 1 1 155.52 mhz 0 1 0 1 32 1 where div_f ref is the divided down reference referred to the 10 mhz to 20 mhz band (see the r section). eference clock (optional) table 9. control register, ctrlb config lol reset misc[4] system reset reset misc[2] d7 d6 d5 d4 d3 d2 d1 d0 0 = lol pin normal operation write a 1 followed by 0 to reset misc[4] write a 1 followed by 0 to reset adn2804 set to 0 write a 1 followed by 0 to reset misc[2] set to 0 set to 0 set to 0 1 = lol pin is static lol table 10. control register, ctrlc config los squelch mode output boost d7 d6 d5 d4 d3 d2 d1 d0 set to 0 set to 0 set to 0 set to 0 set to 0 0 = active high los 0 = squelch data outputs and clock outputs 0 = default output swing 1 = active low los 1 = squelch data outputs or clock outputs 1 = boost output swing
adn2804 rev. b | page 12 of 24 terminology scope probe pin 2.5v vref adn2804 quantizer + ? 10mv p-p vref 05801-013 50? 3k? 50? input sensitivity and input overdrive sensitivity and overdrive specifications for the quantizer involve offset voltage, gain, and noise. the relationship between the logic output of the quantizer and the analog voltage input is shown in figure 12 . for sufficiently large positive input voltage, the output is always logic 1; similarly, for negative inputs, the output is always logic 0. however, the transitions between output logic level 1 and output logic level 0 are not at precisely defined input voltage levels, but occur over a range of input voltages. within this range of input voltages, the output may be either 1 or 0, or it may even fail to attain a valid logic state. the width of this zone is determined by the input voltage noise of the quantizer. the center of the zone is the quantizer input offset voltage. input overdrive is the magnitude of signal required to guarantee the correct logic level with 1 10 ?10 confidence level. figure 13. single-ended sensitivity measurement when the adn2804 is driven differentially (see figure 14 ), sensitivity seems to improve if observing the quantizer input with an oscilloscope probe. this is an illusion caused by the use of a single-ended probe. a 5 mv p-p signal appears to drive the adn2804 quantizer; however, the single-ended probe measures only half the signal. the true quantizer input signal is twice this value, because the other quantizer input is a complementary signal to the signal being observed. noise output input (v p-p) offset overdrive sensitivity (2 overdrive) 1 0 05801-012 scope probe pin 50? 3k ? 2.5v 50 ? vref quantizer + ? nin 5mv p-p v ref 5mv p-p v ref 0 5801-014 figure 12. input sensitivity and input overdrive single-ended vs. differential ac coupling is typically used to drive the inputs to the quantizer. the inputs are internally dc biased to a common- mode potential of ~2.5 v. driving the adn2804 in a single- ended fashion and observing the quantizer input with an oscilloscope probe at the point indicated in figure 14. differential sensitivity measurement figure 13 shows a binary signal with an average value equal to the common-mode potential and instantaneous values both above and below the average value. it is convenient to measure the peak-to-peak amplitude of this signal and call the minimum required value the quantizer sensitivity. referring to los response time los response time is the delay between removal of the input signal and indication of loss of signal (los) at the los output, pin 22. when the inputs are dc-coupled, the los assert time of the ad2804 is 500 ns typical and the deassert time is 400 ns typical. in practice, the time constant produced by the ac coupling at the quantizer input and the 50 on-chip input termination determines the los response time. figure 13 , the sensitivity is twice the overdrive because both positive and negative offsets need to be accommodated. the adn2804 quantizer typically has 3.3 mv p-p sensitivity.
adn2804 rev. b | page 13 of 24 jitter specifications 0.1 acceptable range f c jitter frequency (khz) slope = ?20db/decade jitter gain (db) 0 5801-015 the adn2804 cdr is designed to achieve the best bit- error-rate (ber) performance and to exceed the jitter transfer, generation, and tolerance specifications proposed for sonet/sdh equipment defined in the telcordia technologies specification. jitter is the dynamic displacement of digital signal edges from their long-term average positions, measured in unit intervals (ui), where 1 ui = 1 bit period. jitter on the input data can cause dynamic phase errors on the recovered clock sampling edge. jitter on the recovered clock causes jitter on the retimed data. figure 15. jitter transfer curve the following sections briefly summarize the specifications of jitter generation, transfer, and tolerance in accordance with the telcordia document (gr-253-core, issue 3, september 2000) for the optical interface at the equipment level and the adn2804 performance with respect to those specifications. jitter tolerance the jitter tolerance is defined as the peak-to-peak amplitude of the sinusoidal jitter applied on the input signal, which causes a 1 db power penalty. this is a stress test intended to ensure that no additional penalty is incurred under the operating conditions (see jitter generation figure 16 ). the jitter generation specification limits the amount of jitter that can be generated by the device with no jitter and wander applied at the input. for sonet devices, the jitter generated must be less than 0.01 ui rms and less than 0.1 ui p-p. 15.00 1.50 0.15 f 0 f 1 f 2 f 3 f 4 jitter frequency (khz) slope = ?20db/decade input jitte r amplitude (ui p-p) 0 5801-016 jitter transfer the jitter transfer function is the ratio of the jitter on the output signal to the jitter applied on the input signal vs. the frequency. this parameter measures the amount of jitter on an input signal that can be transferred to the output signal (see figure 15 ). this amount is limited. figure 16. sonet jitter tolerance mask
adn2804 rev. b | page 14 of 24 theory of operation x(s) z(s) recovered clock e(s) input data d/sc psh o/s 1/n d = phase detector gain o = vco gain c = loop integrator psh = phase shifter gain n = divide ratio = 1 cn do s 2 + n psh o s+ 1 z(s) x(s) jitter transfer function = s 2 s 2 d psh c s ++ do cn e(s) x(s) tracking error transfer function 0 5801-017 the adn2804 is a delay- and phase-locked loop circuit for clock recovery and data retiming from an nrz encoded data stream. the phase of the input data signal is tracked by two separate feedback loops, which share a common control voltage. a high speed delay-locked loop path uses a voltage controlled phase shifter to track the high frequency components of input jitter. a separate phase control loop, composed of the vco, tracks the low frequency components of input jitter. the initial frequency of the vco is set by yet a third loop that compares the vco frequency with the input data frequency and sets the coarse tuning voltage. the jitter tracking phase-locked loop controls the vco by the fine-tuning control. figure 17. pll/d ll architecture the delay and phase loops together track the phase of the input data signal. for example, when the clock lags the input data, the phase detector drives the vco to a higher frequency and increases the delay through the phase shifter; both of these actions serve to reduce the phase error between the clock and the data. the faster clock picks up phase, whereas the delayed data loses phase. because the loop filter is an integrator, the static phase error is driven to 0. adn2804 z(s) x(s) frequency (khz) jitter peaking in ordinary pll jitter gain (db) o n psh d psh c 0 5801-018 another view of the circuit is that the phase shifter implements the zero required for frequency compensation of a second-order phase-locked loop, and this zero is placed in the feedback path; therefore, it does not appear in the closed-loop transfer function. jitter peaking in a conventional second-order phase- locked loop is caused by the presence of this zero in the closed- loop transfer function. because this circuit has no zero in the closed-loop transfer, jitter peaking is minimized. figure 18. jitter response vs. conventional pll the delay and phase loops contribute to overall jitter accom- modation. at low frequencies of input jitter on the data signal, the integrator in the loop filter provides high gain to track large jitter amplitudes with small phase error. in this case, the vco is frequency modulated, and jitter is tracked as in an ordinary phase-locked loop. the amount of low frequency jitter that can be tracked is a function of the vco tuning range. a wider tuning range gives larger accommodation of low frequency jitter. the internal loop control voltage remains small for small phase errors; therefore, the phase shifter remains close to the center of its range and thus contributes little to the low frequency jitter accommodation. the delay and phase loops together simultaneously provide wideband jitter accommodation and narrow-band jitter filtering. the linearized block diagram in figure 17 shows that the jitter transfer function, z(s)/x(s), provides excellent second- order low-pass filtering. note that the jitter transfer has no zero, unlike an ordinary second-order phase-locked loop. this means that the main pll loop has virtually no jitter peaking (see figure 18 ), making this circuit ideal for signal regenerator applications, where jitter peaking in a cascade of regenerators can contribute to hazardous jitter accumulation. the error transfer, e(s)/x(s), has the same high-pass form as an ordinary phase-locked loop. this transfer function can be optimized to accommodate a significant amount of wideband jitter, because the jitter transfer function, z(s)/x(s), provides the narrow-band jitter filtering.
adn2804 rev. b | page 15 of 24 at medium jitter frequencies, the gain and tuning range of the vco are not large enough to track input jitter. in this case, the vco control voltage becomes large and saturates, and the vco frequency dwells at one extreme of its tuning range. the size of the vco tuning range, therefore, has only a small effect on the jitter accommodation. the delay-locked loop control voltage is now larger; therefore, the phase shifter takes on the burden of tracking the input jitter. the phase shifter range, in ui, can be seen as a broad plateau on the jitter tolerance curve. the phase shifter has a minimum range of 2 ui at all data rates. the gain of the loop integrator is small for high jitter frequencies; therefore, larger phase differences are needed to increase the loop control voltage enough to tune the range of the phase shifter. however, large phase errors at high jitter frequencies cannot be tolerated. in this region, the gain of the integrator determines the jitter accommodation. because the gain of the loop integrator declines linearly with frequency, jitter accommodation is lower with higher jitter frequency. at the highest frequencies, the loop gain is very small, and little tuning of the phase shifter can be expected. in this case, jitter accommodation is determined by the eye opening of the input data, the static phase error, and the residual loop jitter generation. the jitter accommodation is roughly 0.5 ui in this region. the corner frequency between the declining slope and the flat region is the closed-loop bandwidth of the delay-locked loop, which is roughly 1.0 mhz at 622 mbps.
adn2804 rev. b | page 16 of 24 functional description loss-of-signal (los) detector frequency acquisition the receiver front-end los detector circuit detects when the input signal level falls below a user-adjustable threshold. the threshold is set with a single external resistor from pin 9, thradj, to vee. the los comparator trip point vs. the resistor value is shown in the adn2804 acquires frequency from the data. the lock detector circuit compares the frequency of the vco and the frequency of the incoming data. when these frequencies differ by more than 1000 ppm, lol is asserted. this initiates a frequency acquisition cycle. when the vco frequency is within 250 ppm of the data frequency, lol is deasserted. figure 6 . if the input level to the adn2804 drops below the programmed los threshold, the output of the los detector, los (pin 22), is asserted to logic 1. the los detectors response time is ~500 ns by design, but is dominated by the rc time constant in ac-coupled applications. the los pin defaults to active high. however, setting bit ctrlc[2] to 1, configures the los pin as active low. once lol is deasserted, the frequency-locked loop is turned off. the pll/dll pulls the vco frequency in the rest of the way until the vco frequency equals the data frequency. the frequency loop requires a single external capacitor between cf1 and cf2, pin 14 and pin 15. a 0.47 f 20%, x7r ceramic chip capacitor with <10 na leakage current is recommended. leakage current of the capacitor can be calculated by dividing the maximum voltage across the 0.47 f capacitor, ~3 v, by the insulation resistance of the capacitor. the insulation resistance of the 0.47 f capacitor should be greater than 300 m. there is typically 6 db of electrical hysteresis designed into the los detector to prevent chatter on the los pin. if the input level drops below the programmed los threshold causing the los pin to assert, the los pin deasserts after the input level increases to 6 db (2) above the los threshold (see figure 19 ). hysteresis los output input level los threshold t input voltage (v diff ) 0 5801-019 limiting amplifier the limiting amplifier has differential inputs (pin/nin) that are internally terminated with 50 to an on-chip voltage reference (vref = 2.5 v typically). the inputs are typically ac-coupled externally, although dc coupling is possible as long as the input common-mode voltage remains above 2.5 v (see figure 27 to figure 29 in the applications information section). input offset is factory trimmed to achieve better than 3.3 mv typical sensitivity with minimal drift. the limiting amplifier can be driven differentially or in a single-ended fashion. figure 19. los detector hysteresis slice adjust the los detector and the slice level adjust can be used simultaneously on the adn2804. this means that any offset added to the input signal by the slice adjust pins does not affect the los detectors measurement of the absolute input level. the quantizer slicing level can be offset by 100 mv to mitigate the effect of amplified spontaneous emission (ase) noise or duty cycle distortion by applying a differential voltage input of up to 0.95 v to the slicep and slicen inputs. if no adjustment of the slice level is needed, slicep and slicen should be tied to vee. the gain of the slice adjustment is ~0.11 v/v.
adn2804 rev. b | page 17 of 24 static lol mode lock detector operation the adn2804 implements a static lol feature that indicates if a loss-of-lock condition has ever occurred. this feature remains asserted, even if the adn2804 regains lock, until the static lol bit is manually reset. the i 2 c register bit, misc[4], is the static lol bit. if there is ever an occurrence of a loss-of-lock condition, this bit is internally asserted to logic high. the misc[4] bit remains high even after the adn2804 has reacquired lock to a new data rate. this bit can be reset by writing a 1 followed by 0 to i 2 c register bit ctrlb[6]. once reset, the misc[4] bit remains deasserted until another loss-of-lock condition occurs. the lock detector on the adn2804 has three modes of operation: normal mode, refclk mode, and static lol mode. normal mode in normal mode, the adn2804 is a cdr that locks onto a 622 mbps data rate without the use of a reference clock as an acquisition aid. in this mode, the lock detector monitors the frequency difference between the vco and the input data frequency and deasserts the loss of lock signal, which appears on pin 16, lol, when the vco is within 250 ppm of the data frequency. this enables the d/pll, which pulls the vco frequency in the remaining amount and acquires phase lock. once locked, if the input frequency error exceeds 1000 ppm (0.1%), the loss-of-lock signal is reasserted and control returns to the frequency loop, which begins a new frequency acquisition. the lol pin remains asserted until the vco locks onto a valid input data stream to within 250 ppm frequency error. this hysteresis is shown in wr it ing a 1 to i 2 c register bit ctrlb[7] causes the lol pin, pin 16, to become a static lol indicator. in this mode, the lol pin mirrors the contents of the misc[4] bit and has the functionality described in the previous paragraph. the ctrlb[7] bit defaults to 0. in this mode, the lol pin operates in the normal operating mode, that is, it is asserted only when the adn2804 is in acquisition mode and deasserts when the adn2804 has reacquired lock. figure 20 . lol 0 ?250 250 1000 f vco error (ppm) ?1000 1 0 5801-020 squelch modes two modes for the squelch pin are available with the adn2804: squelch data outputs and clock outputs mode and squelch data outputs or clock outputs mode. squelch data outputs and clock outputs mode is selected when ctrlc[1] is 0 (default mode). in this mode, when the squelch input, pin 27, is driven to a ttl high state, both the data outputs (dataoutn and dataoutp) and the clock outputs (clkoutn and clkoutp) are set to the zero state to suppress downstream processing. if the squelch function is not required, pin 27 should be tied to vee. figure 20. transfer function of lol lol detector operation using a reference clock in refclk mode, a reference clock is used as an acquisition aid to lock the adn2804 vco. lock-to-reference mode is enabled by setting ctrla[0] to 1. the user also needs to write to the ctrla[7, 6] and ctrla[5:2] bits to set the reference frequency range and the divide ratio of the data rate with respect to the reference frequency. for more details, see the squelch data outputs or clock outputs mode is selected when ctrlc[1] is 1. in this mode, when the squelch input is driven to a high state, the dataoutn and dataoutp pins are squelched. when the squelch input is driven to a low state, the clkoutn and clkoutp pins are squelched. this is especially useful in repeater applications, where the recovered clock may not be needed. reference clock (optional) section. in this mode, the lock detector monitors the difference in frequency between the divided down vco and the divided down reference clock. the loss-of-lock signal, which appears on pin 16, lol, is deasserted when the vco is within 250 ppm of the desired frequency. this enables the d/pll, which pulls the vco frequency in the remaining amount with respect to the input data and acquires phase lock. once locked, if the input frequency error exceeds 1000 ppm (0.1%), the loss-of-lock signal is reasserted and control returns to the frequency loop, which reacquires with respect to the reference clock. the lol pin remains asserted until the vco frequency is within 250 ppm of the desired frequency. this hysteresis is shown in i 2 c interface the adn2804 supports a 2-wire, i 2 c-compatible serial bus driving multiple peripherals. two inputs, serial data (sda) and serial clock (sck), carry information to and from any device connected to the bus. each slave device is recognized by a unique address. the adn2804 has two possible 7-bit slave addresses for both read and write operations. the msb of the 7-bit slave address is factory programmed to 1. b5 of the slave address is set by pin 19, saddr5. slave address bits [4:0] are defaulted to all 0s. the slave address consists of the seven msbs of an 8-bit word. the lsb of the word either sets a read or write operation (see figure 20 . figure 7 ). logic 1 corresponds to a read operation, while logic 0 corresponds to a write operation.
adn2804 rev. b | page 18 of 24 to control the device on the bus, the following protocol must be followed. first, the master initiates a data transfer by establish- ing a start condition, defined by a high-to-low transition on sda while sck remains high. this indicates that an address/ data stream follows. all peripherals respond to the start condition and shift the next eight bits (the 7-bit address and the r/w bit). the bits are transferred from msb to lsb. the peripheral that recognizes the transmitted address responds by pulling the data line low during the ninth clock pulse. this is known as an acknowledge bit. all other devices withdraw from the bus at this point and maintain an idle condition. the idle condition is where the device monitors the sda and sck lines, waiting for the start condition and correct transmitted address. the r/w bit determines the direction of the data. logic 0 on the lsb of the first byte means that the master writes information to the peripheral. logic 1 on the lsb of the first byte means that the master reads information from the peripheral. the adn2804 acts as a standard slave device on the bus. the data on the sda pin is eight bits long, supporting the 7-bit addresses plus the r/w bit. the adn2804 has eight subaddresses to enable the user-accessible internal registers (see table 6 through table 10). it, therefore, interprets the first byte as the device address and the second byte as the starting subaddress. auto-increment mode is supported, allowing data to be read from or written to the starting subaddress and each subsequent address without manually addressing the subsequent subaddress. a data transfer is always terminated by a stop condition. the user can also access any unique subaddress register on a one-by-one basis without updating all registers. stop and start conditions can be detected at any stage of the data transfer. if these conditions are asserted out of sequence with normal read and write operations, they cause an immediate jump to the idle condition. during a given sck high period, the user should issue one start condition, one stop condition, or a single stop condition followed by a single start condition. if an invalid subaddress is issued by the user, the adn2804 does not issue an acknowledge and returns to the idle condition. if the user exceeds the highest subaddress while reading back in auto- increment mode, then the highest subaddress register contents continue to be output until the master device issues a no acknow- ledge. this indicates the end of a read. in a no-acknowledge condition, the sdata line is not pulled low on the ninth pulse. see figure 8 and figure 9 for sample write and read data transfers and figure 10 for a more detailed timing diagram. additional features available via the i 2 c interface los configuration the los detector output, pin 22, can be configured to be either active high or active low. if ctrlc[2] is set to logic 0 (default), the los pin is active high when a loss-of-signal condition is detected. writing a 1 to ctrlc[2] configures the los pin to be active low when a loss-of-signal condition is detected. system reset a frequency acquisition can be initiated by writing a 1 followed by a 0 to the i 2 c register bit ctrlb[5]. this initiates a new frequency acquisition while keeping the adn2804 in its previously programmed operating mode, as set in registers ctrl[a], ctrl[b], and ctrl[c].
adn2804 rev. b | page 19 of 24 reference clock (optional) a reference clock is not required to perform clock and data recovery with the adn2804; however, support for an optional reference clock is provided. the reference clock can be driven differentially or in a single-ended fashion. if the reference clock is not being used, refclkp should be tied to vcc, and refclkn can be left floating or tied to vee (the inputs are internally terminated to vcc/2). see figure 21 through figure 23 for sample configurations. the refclk input buffer accepts any differential signal with a peak-to-peak differential amplitude of greater than 100 mv (for example, lvpecl or lvds) or a standard single-ended, low voltage ttl input, providing maximum system flexibility. phase noise and duty cycle of the reference clock are not critical, and 100 ppm accuracy is sufficient. 100k? vcc/2 100k? adn2804 refclkp 10 11 refclkn buffer 0 5801-021 figure 21. differential refclk configuration 100k ? vcc/2 100k ? adn2804 refclkp out refclkn buffer v cc clk osc 0 5801-022 figure 22. single-ended refclk configuration 100k ? vcc/2 100k ? adn2804 refclkp 10 11 nc refclkn buffer vcc 0 5801-023 figure 23. no refc lk configuration there are two mutually exclusive uses, or modes, of the reference clock. the reference clock can be used either to help the adn2804 lock onto data or to measure the frequency of the incoming data to within 0.01%. the modes are mutually exclusive because in the first use the user knows exactly what the data rate is and wants to force the part to lock onto only that data rate, and in the second use the user does not know what the data rate is and wants to measure it. lock-to-reference mode is enabled by writing a 1 to i 2 c register bit ctrla[0]. fine data rate readback mode is enabled by writing a 1 to i 2 c register bit ctrla[1]. writing a 1 to both of these bits at the same time causes an indeterminate state and is not supported. using the reference clock to lock onto data in this mode, the adn2804 locks onto a frequency derived from the reference clock according to data rate /2 ctrla[5:2] = refclk /2 ctrla[7, 6] the user must provide a reference clock that is a function of the data rate. by default, the adn2804 expects a reference clock of 19.44 mhz. other options are 38.88 mhz, 77.76 mhz, and 155.52 mhz, which are selected by programming ctrla[7, 6]. ctrla[5:2] should be programmed to [0101] for all cases. table 11. ctrla settings ctrla[7, 6] range (mhz) ctrla[5:2] ratio 00 19.44 0101 2 5 01 38.88 0101 2 5 10 77.76 0101 2 5 11 155.52 0101 2 5 for example, if the reference clock frequency is 38.88 mhz and the input data rate is 622.08 mbps, ctrla[7, 6] is set to [01] to produce a divided-down reference clock of 19.44 mhz, and ctrla[5:2] is set to [0101], that is, 5, because 622.08 mbps/19.44 mhz = 2 5 in this mode, if the adn2804 loses lock for any reason, it relocks onto the reference clock and continues to output a stable clock. while the adn2804 is operating in lock-to-reference mode, a 0 to 1 transition should be written into the ctrla[0] bit to initiate a lock-to-reference clock command.
adn2804 rev. b | page 20 of 24 3. read back misc[2]. if it is 0, the measurement is not complete. if it is 1, the measurement is complete and the data rate can be read back on freq[22:0]. the time for a data rate measurement is typically 80 ms. using the reference clock to measure data frequency the user can also provide a reference clock to measure the recovered data frequency. in this case, the user provides a reference clock, and the adn2804 compares the frequency of the incoming data to the incoming reference clock and returns a ratio of the two frequencies to within 0.01% (100 ppm) accuracy. the accuracy error of the reference clock is added to the accuracy of the adn2804 data rate measurement. for example, if a 100 ppm accuracy reference clock is used, the total accuracy of the measure- ment is within 200 ppm. 4. read back the data rate from freq2[6:0], freq1[7:0], and freq0[7:0]. the data rate can be determined by [] ( ) )_14( 2/ 0.22 ratesel refclk datarate f freq f + = where: freq[22:0] is the reading from freq2[6:0] (msb byte, freq1[7:0], and freq0[7:0] (lsb byte). f datarate is the data rate (mbps). f refclk is the refclk frequency (mhz). sel_rate is the setting from ctrla[7, 6]. the reference clock can range from 10 mhz to 160 mhz. by default, the adn2804 expects a reference clock between 10 mhz and 20 mhz. if the reference clock is between 20 mhz and 40 mhz, 40 mhz and 80 mhz, or 80 mhz and 160 mhz, the user must configure the adn2804 for the correct reference frequency range by setting two bits of the ctrla register, ctrla[7, 6]. using the reference clock to determine the frequency of the incoming data does not affect the manner in which the part locks onto data. in this mode, the reference clock is used only to determine the frequency of the data. for example, if the reference clock frequency is 32 mhz, sel_rate = 1, because the reference frequency falls into the 20 mhz to 40 mhz range, setting ctrla[7, 6] to [01],. assume for this example that the input data rate is 622.08 mb/s (oc12). after following step 1 through step 4, the value that is read back on freq[22:0] = 0x9b851, which is equal to 637 10 3 . plugging this value into the equation yields prior to reading back the data rate using the reference clock, the ctrla[7, 6] bits must be set to the appropriate frequency range with respect to the reference clock being used. a fine data rate readback is then executed as follows: 637e3 32e6/2 (14 + 1) = 622.08 mbps if subsequent frequency measurements are required, ctrla[1] should remain set to 1. it does not need to be reset. the measurement process is reset by writing a 1 followed by a 0 to ctrlb[3]. this initiates a new data rate measurement. follow step 2 through step 4 to read back the new data rate. 1. write a 1 to ctrla[1]. this enables the fine data rate measurement capability of the adn2804. this bit is level sensitive and can perform subsequent frequency measurements without being reset. 2. reset misc[2] by writing a 1 followed by a 0 to ctrlb[3]. this initiates a new data rate measurement. note that a data rate readback is valid only if lol is low. if lol is high, the data rate readback is invalid. table 12. d22 d21 ... d17 d16 d15 d14 ... d9 d8 d7 d6 ... d1 d0 freq2[6:0] freq1[7:0] freq0[7:0]
adn2804 rev. b | page 21 of 24 applications information pcb design guidelines proper rf pcb design techniques must be used for optimal performance. power supply connections and ground planes use of one low impedance ground plane is recommended. the vee pins should be soldered directly to the ground plane to reduce series inductance. if the ground plane is an internal plane and connections to the ground plane are made through vias, multiple vias can be used in parallel to reduce the series inductance, especially on pin 23, which is the ground return for the output buffers. the exposed pad should be connected to the gnd plane using plugged vias so that solder does not leak through the vias during reflow. use of a 22 f electrolytic capacitor between vcc and vee is recommended at the location where the 3.3 v supply enters the pcb. when using 0.1 f and 1 nf ceramic chip capacitors, they should be placed between adn2804 supply pins vcc and vee, as close as possible to the adn2804 vcc pins. if connections to the supply and ground are made through vias, the use of multiple vias in parallel helps to reduce series inductance, especially on pin 24, which supplies power to the high speed clkoutp/clkoutn and dataoutp/ dataoutn output buffers. refer to figure 24 for the recommended connections. by placing the power supply and gnd planes adjacent to each other and using close spacing between the planes, excellent high frequency decoupling can be realized. the capacitance is given by ? ? pf/0.88 r da c plane ? where: ? r is the dielectric constant of the pcb material. a is the area of the overlap of power and gnd planes (cm 2 ). d is the separation between planes (mm). for fr-4, ? r = 4.4 and d = 0.25 mm; therefore, c plane ~ 15.5a (pf) c i 2 c controller i 2 c controller los sda sck saddr5 50? transmission lines dataoutp dataoutn clkoutp clkoutn 0.1f 22f 1nf 0.1f 0.1f 0.1f 1.6f 1.6f 0.1f 0.47f 20% >300m ? insulation resistance 1nf 1nf 1nf 0.1f 1nf + v cc 50? 50? tia r th vcc nc c vcc vcc 1 vcc 2 vcc 3 vref 4 nin 5 pin 6 slicep 7 slicen 8 vee 24 vcc 23 vee 22 21 20 19 18 vcc 17 vee 9 thradj 10 refclkp 11 refclkn 12 vcc 13 vee 14 cf2 15 cf1 16 lol 32 vcc 31 vcc 30 vee 29 dataoutp 28 dataoutn 27 squelch 26 clkoutp 25 clkoutn exposed pad tied off to vee plane with vias 05801-031 figure 24. typical adn2804 applications circuit
adn2804 rev. b | page 22 of 24 choosing ac coupling capacitors transmission lines ac coupling capacitors at the input (pin, nin) and output (dataoutp, dataoutn) of the adn2804 can be optimized for the application. when choosing the capacitors, the time constant formed with the two 50 resistors in the signal path must be considered. when a large number of consecutive identical digits (cids) are applied, the capacitor voltage can droop due to baseline wander (see minimizing reflections in th e adn2804 requires use of 50 transmission lines for all pins with high frequency input and output signals, including pin, nin, clkoutp, clkoutn, dataoutp, and dataoutn (also refclkp and refclkn, if a high frequency reference clock is used, such as 155 mhz). it is also necessary for the pin/nin input traces to be matched in length and for the clkoutp/clkoutn and dataoutp/dataoutn output traces to be matched in length to avoid skew between the differential traces. figure 26 ), causing pattern- dependent jitter (pdj). the user must determine how much droop is tolerable and choose an ac coupling capacitor based on that amount of droop. the amount of pdj can then be approximated based on the capacitor selection. the actual capacitor value selection can require some trade-offs between droop and pdj. the high speed inputs, pin and nin, are internally terminated with 50 to an internal reference voltage (see figure 25 ). a 0.1 f is recommended between vref, pin 3, and gnd to provide an ac ground for the inputs. for example, assuming that 2% droop can be tolerated, the maximum differential droop is 4%. normalizing to v p-p: as with any high speed, mixed-signal design, take care to keep all high speed digital traces away from sensitive analog nodes. droop = v = 0.04 v = 0.5 v p-p (1 ? e ?t/ ); therefore, = 12t c in c in 0.1f nin pin adn2804 2.5v vref tia v cc 0 5801-026 50? 50? 3 ? where: is the rc time constant (c is the ac coupling capacitor, r = 100 seen by c). t is the total discharge time, which is equal to nt , where n is the number of cids, and t is the bit period. the capacitor value can then be calculated by combining the equations for and t: c = 12 nt / r figure 25. adn2804 ac-coupled input configuration soldering guidelines for lead frame chip scale package once the capacitor value is selected, the pdj can be approximated as the lands on the 32-lead lfcsp are rectangular. the printed circuit board (pcb) pad for these should be 0.1 mm longer than the package land length and 0.05 mm wider than the package land width. the land should be centered on the pad. this ensures that the solder joint size is maximized. the bottom of the chip scale package has a central exposed pad. the pad on the pcb should be at least as large as this exposed pad. the user must connect the exposed pad to vee using plugged vias so that solder does not leak through the vias during reflow. this ensures a solid connection from the exposed pad to vee. pdj pspp = 0.5 t r (1 ? e (? nt / rc ) )/0.6 where: pdj pspp is the amount of pattern-dependent jitter allowed (<0.01 ui p-p typical). t r is the rise time, which is equal to 0.22/ bw , where bw ~ 0.7 (bit rate). note that this expression for t r is accurate only for the inputs. the output rise time for the adn2804 is ~100 ps regardless of the data rate.
adn2804 rev. b | page 23 of 24 50? 50? pin v ref nin c in c out c out v1 c in v1b v2 v2b limamp cdr + ? dataoutp dataoutn 1 v1 v1b v2 v2b v diff 234 vref vth adn2804 v diff = v2?v2b vth = adn2804 quantizer threshold notes: 1. during data patterns with high transition density, differential dc voltage at v1 and v2 is zero. 2. when the output of the tia goes to cid, v1 and v1b are driven to different dc levels. v2 and v2b discharge to the vref level, which effectively introduces a differential dc offset across the ac coupling capacitors. 3. when the burst of data starts again, the differential dc offset across the ac coupling capacitors is applied to the input levels causing a dc shift in the differential input. this shift is large enough such that one of the states, either high or low depending on the levels of v1and v1b when the tia went to cid, is canceled out. the quantizer does not recognize this as a valid state. 4. the dc offset slowly discharges until the differential input voltage exceeds the sensitivity of the adn2804. the quantizer can recognize both high and low states at this point. 0 5801-027 tia v cc figure 26. example of baseline wander pin input (v) v p-p = pin ? nin = 2 v se = 10mv at sensitivity v se = 5mv min v cm = 2.3v min (dc-coupled) nin 0 5801-029 dc-coupled application the inputs to the adn2804 can also be dc-coupled. this may be necessary in burst mode applications, where there are long periods of cids, and baseline wander cannot be tolerated. if the inputs to the adn2804 are dc-coupled, care must be taken not to violate the input range and common-mode level require- ments of the adn2804 (see figure 27 through figure 29 ). if dc coupling is required and the output levels of the tia do not adhere to the levels shown in figure 28 , level shifting must be performed and/or an attenuator must be placed between the tia outputs and the adn2804 inputs. figure 28. minimum allowed dc-coupled input levels pin input (v) v p-p = pin ? nin = 2 v se = 2.0v max v se = 1.0v max v cm = 2.3v (dc-coupled) nin 0 5801-030 0.1f nin pin adn2804 2.5v vref tia v cc 05801-028 50 ? 50 ? 50? 50? 3k ? figure 27. dc-cou pled application figure 29. maximum allowed dc-coupled input levels
adn2804 rev. b | page 24 of 24 outline dimensions compliant to jedec standards mo-220-vhhd-2 011708-a 0.30 0.23 0.18 0.20 ref 0.80 max 0.65 typ 0.05 max 0.02 nom 12 max 1.00 0.85 0.80 seating plane coplanarity 0.08 1 32 8 9 25 24 16 17 0.50 0.40 0.30 3.50 ref 0.50 bsc pin 1 indi c ator top view 5.00 bsc sq 4.75 bsc sq 3.25 3.10 sq 2.95 pin 1 indicator 0.60 max 0.60 max 0.25 min exposed pad (bottom view) for proper connection of the exposed pad, refer to the pin configuration and function descriptions section of this data sheet. figure 30. 32-lead lead frame chip scale package [lfcsp_vq] 5 mm 5 mm body, very thin quad (cp-32-2) dimensions shown in millimeters ordering guide model 1 temperature range package description package option adn2804acpz ?40c to +85c 32-lead lfcsp_vq cp-32-2 adn2804acpz-500rl7 ?40c to +85c 32-lead lfcsp_vq, tape-reel, 500 pieces cp-32-2 ADN2804ACPZ-RL7 ?40c to +85c 32-lead lfcsp_vq, tape-reel, 1500 pieces cp-32-2 eval-adn2804ebz evaluation board 1 z = rohs compliant part. i 2 c refers to a communications protocol originally developed by philips semiconductors (now nxp semiconductors). ?2006C2010 analog devices, inc. all rights reserved. trademarks and registered trademarks are the prop erty of their respective owners. d05801-0-5/10(b)

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